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MIC2124
Constant Frequency, Synchronous Current Mode Buck Controller Hyper Speed ControlTM Family
General Description
The Micrel MIC2124 is a member of the Hyper Speed ControTMl family of DC-DC controllers. It uses an adaptive on-time current mode control scheme and operates at a constant frequency. The MIC2124 operates over a supply range of +3V to +18V, and is independent of the IC supply voltage VIN. It operates at a fixed 300kHz switching frequency and can be used to provide up to 25A of output current. The output voltage is adjustable down to +0.8V. The MIC2124 includes an EN/COMP pin that can be pulled low to shut down the converter. The MIC2124 optimizes performance and ensures stability with external compensation. The UVLO is provided to ensure proper operation under power-sag conditions and to make sure that the external power MOSFET has enough voltage to work with. An internal digital soft-start ensures reduced inrush current. Cycle-by-cycle current limiting ensures FET protection. The MIC2124 is available in a 10-pin MSOP package with a junction temperature operating range from -40C to +125C.
Features
* * * * * * * * * * * * * * * +3V to +18V input voltage 25A output current capability TM Any Capacitor stable - Zero ESR to high ESR Output down to 0.8V with 1% FB accuracy Up to 94% efficiency 300kHz switching frequency All N-Channel MOSFET design Shutdown feature with EN/COMP No current-sense resistor needed Internal 4ms digital soft-start Thermal shutdown Pre-bias output safe Cycle-by-Cycle foldback current-limit protection 10-pin MSOP package -40C to +125C junction temperature range
Applications
* * * * * * * Printers and scanners Graphic and video cards PCs and servers Microprocessor core supply Low-voltage distributed power Telecommunication and networking Set-top box, gateways and router
Typical Application
12V to 3.3V Efficiency
100
90 EFFICIENCY (%)
80
70
VHSD = 12V VIN = 5V
0 2 4 6 8 O UT PUT CURRENT (A) 10
60
MIC2124 Adjustable Output 300kHz Buck Converter
Hyper Speed Control and Any Capacitor are trademarks of Micrel, Inc.
Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
June 2010
M9999-060810-D
Micrel, Inc.
MIC2124
Ordering Information
Part Number MIC2124YMM Voltage Adj. Switching Frequency 300kHz Junction Temp. Range -40 to +125C Package 10-Pin MSOP Lead Finish Pb-Free
Pin Configuration
10-Pin MSOP (MM)
Pin Description
Pin Number 1 Pin Name HSD Pin Function High-Side N-MOSFET Drain Connection (Input): Input voltage for the internal sensing of external power stage supply. The HSD operating voltage range is from 3V to 18V. Input capacitors between HSD and the power ground (PGND) are required. Enable (Input): Floating = enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically 1mA). COMP (Output): Output of the gm error amplifier and connects to the components for the external compensation. Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. Signal ground. GND is the ground path for the device input voltage VIN and the control circuitry. The loop for the signal ground should be separate from the power ground (PGND) loop. Input Voltage (Input): Power to the internal reference and control sections of the MIC2124. The IN operating voltage range is from 3V to 5.5V. A 2.2F ceramic capacitors from IN to GND are recommended for clean operation. Connect IN to HSD when VHSD < 5.5V. Low-Side Drive (Output): High-current driver output for external low-side MOSFET. The DL driving voltage swings from ground-to-IN. Power Ground. PGND is the ground path for the MIC2124 buck converter power stage. The PGND pin connects to the sources of low-side N-Channel MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the Signal ground (GND) loop. High-Side Drive (Output): High-current driver output for external high-side MOSFET. The DH driving voltage is floating on the switch node voltage (LX). It swings from VLX to VBST. Switch Node (Input): High-current output driver return. The LX pin connects directly to the switch node. Due to the high speed switching on this pin, the LX pin should be routed away from sensitive nodes. Current Sense input (Input): LX pin also senses the current for the current mode control and short circuit protection by monitoring the voltage across the low-side MOSFET during OFF time. In order to sense the current accurately, connect the low-side MOSFET drain to LX using a Kelvin connection. Boost (Output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is connected between the IN pin and the BST pin. A boost capacitor of 0.1F is connected between the BST pin and the LX pin. Adding a small resistor at BST pin can slow down the turn-on time of highside N-Channel MOSFETs.
2
EN/COMP
3
FB
4 5
GND IN
6 7
DL PGND
8 9
DH LX
10
BST
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Absolute Maximum Ratings(1)
IN, FB to GND .................................................. -0.3V to +6V BST to LX ......................................................... -0.3V to +6V BST to GND ................................................... -0.3V to +35V DH to LX.............................................-0.3V to (VBST + 0.3V) DL, COMP to GND ...............................-0.3V to (VIN + 0.3V) HSD to GND..................................................... -0.3V to 29V PGND to GND ............................................... -0.3V to +0.3V (3) Power Dissipation TA=70C ................... Internally Limited Storage Temperature (TS)..........................-65C to +150C Lead Temperature (soldering, 10sec) ........................ 260C
Operating Ratings(2)
Input Voltage (VIN) ............................................ 3.0V to 5.5V Supply Voltage (VHSD) ....................................... 3.0V to 18V Ambient Temperature (TA) ...........................-40C to +85C Junction Temperature (TJ) .........................-40C to +125C Junction Thermal Resistance MSOP (JA) ............130C/W Junction Thermal Resistance MSOP (JC) ..............43C/W
Electrical Characteristics(5)
VHSD = 13.2V, VIN = 5V, VBST - VLX = 5V; TA = 25C, unless noted. Bold values indicate -40C TA +85C.
Parameter General Operating Input Voltage (VIN) HSD Voltage Range (VHSD) Quiescent Supply Current Shutdown Current Under-Voltage Lockout Under-voltage Lockout Trip Level UVLO Hysteresis DC-DC Controller Output-Voltage Adjust Range (VOUT) Error Amplifier FB Regulation Voltage FB Regulation Voltage Transconductance gm COMP Output Voltage Swing FB Input Leakage Current On Timer Switching Frequency Maximum Duty Cycle Minimum Duty Cycle Short Current Protection Current Limit 1 Current Limit 2 VFB = 0.8V VFB = 0V 110 21 127 36 145 51 mV mV VHSD = 4V, VIN = 5V, VLX = 3.33V FB > 0.8V 240 89 300 91 0 360 93 kHz % % VFB = 0.8V 0C TA 85C -40C TA 85C -1 -2.5 70 0.5 1 0.2 0.2 110 1 1 160 2.3 500 % % S V nA Depends on external components and the maximum duty cycle. 0.8 V Rising edge 2.5 2.7 40 2.93 V mV VFB = 1.5V VEN/COMP = GND
(6)
Conditions
Min 3.0 3.0
Typ
Max 5.5 18
Units V V mA mA
1.4 1
2 2
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Parameter FET Drivers DH, DL Output Low Voltage DH, DL Output High Voltage DH On-Resistance, High State DH On-Resistance, Low State DL On-Resistance, High State DL On-Resistance, Low State LX, BST, HSD Leakage Current Thermal Protection Over-temperature Shutdown Over-temperature Shutdown Hysteresis Shutdown Control EN/COMP Logic Level High EN/COMP Logic Level Low EN/COMP Hysteresis EN/COMP Pull-up Current
Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating.
MIC2124
Conditions ISINK = 10mA ISOURCE = 10mA, measured the difference between VBST-VDH, VIN-VDL 0.1 2 1.5 2 1 TA = 25C 160 5 3 3 3 2 30 Min Typ Max 0.1 Units V V A C C
3V < VIN <5.5V 3V < VIN <5.5V 3V < VIN <5.5V VEN/COMP = 0V
0.5
0.4 0.4 26 47 100 0.25
V V mV A
3. The maximum allowable power dissipation of any TA is PD(max) = (TJ(max)-TA) / JA. Exceeding the maximum allowable power dissipation will result in excessive die temperature, and the regulator will go into thermal shutdown. 4. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 5. Specification for packaged product only. 6. The application is fully functional at low IN (supply of the control section) if the external MOSFETs have enough low voltage VTH.
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Typical Characteristics
Efficiency vs. Load Current
100
Output Voltage vs. Load Current
2.60
Output Voltage Change vs. Input Voltage
70 OUTPUT VOTLAGE CHANGE(mV) 50
VOUT = 2.5V
90 80 70
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
2.55
VOUT = 2.5V
30 10 -10 -30 -50 -70
VOUT = 1.5V
60 50 40 0.1 1.0 LOAD CURRENT (A) 10.0
2.50
VIN = V HSD = 5V
2.45
VOUT = 0.8V VIN = VHSD IOUT = 5A
V IN = V HSD = 5V
2.40 0 1 2 3 LOAD CURRENT (A) 4 5
3.0
3.5
4.0 4.5 5.0 INPUT VOTLAGE (V)
5.5
Output Voltage vs. Input Voltage
0.810
Switching Frequency vs. Input Voltage
350 SWITCHING FREQUENCY (kHz)
OUTPUT VOTLAGE (V)
0.805
330
310
-40C
25C
0.800
290
0.795
270
85C
VIN = VHSD VOUT = 1.8V IOUT = 2.5A
IOUT = 5A
0.790 10.8
250
11.4 12.0 12.6 INPUT VOTLAGE (V) 13.2
3.0
3.5
4.0 4.5 5.0 INPUT VOTLAGE (V)
5.5
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Functional Characteristics
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Typical Characteristics
Feedback Voltage v s. Temperature
0.805 0.804
SWITCHING FRE QUENCY (kHz)
Switching Frequency v s. Load
360
SWITCHING FRE QUENCY (kHz)
Switching Frequency v s. HSD Voltage
360 340 320 300 280 260 240
VIN = 5V VOUT = 1.8V IOUT = 5A
FEEDBACK VOLTAG E (V)
0.803 0.802 0.801 0.800 0.799 0.798 0.797 0.796 0.795 -40 -20 0 20 40 60 80 100 120 T EM PERAT URE (C)
VIN = 5V
340 320 300 280 260 240 0 2 4 6 8 OUT PUT CURRENT (A) 10
VHSD = 12V VIN = 5V
3
6
9 12 15 VHSD VO LT AGE (V)
18
Switching Frequency v s. Tem perature
360
SWITCHING FREQUENCY (kHz)
Current-Limit Threshold vs. Feedback Voltage
CURRENT-LIMIT THRESHOLD (mV)
Current Limit Threshold v s. Temperature
150 120 90 60 30 0
FB = 0V FB = 0.8V
150 120 90 60
VIN= 5V
CURRENT LIMIT THRESHOLD (m V)
340 320 300 280 260 240 -40 -20 0 20 40 60 80 100 120 T EMPERAT URE (C)
VHSD = 5V VIN = 5V
30 0 0 20 40 60 80 FEEDBACK VOTLAGE (%) 100
-40
-20
0 20 40 60 80 T EM PERAT URE (C)
100 120
Shutdown Current v s. Input Voltage
1000 SHUTDO WN CURRENT (A) 800 600 400 200 0 3 3.5 4 4.5 5 INPUT VO LT AG E (V) 5.5
VHSD = 12V
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Functional Diagram
Figure 1. MIC2124 Block Diagram
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Functional Description
The MIC2124 is an adaptive on-time current mode synchronous buck controller built for low cost and high performance. It is designed for wide input voltage range from 3V to 18V and for high output power buck converters. An estimated-ON-time method is applied in MIC2124 to obtain a constant switching frequency and to simplify the control compensation. The over-current protection is implemented without the use of an external sense resistor. It includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time. Theory of Operation The MIC2124 is an adaptive on-time current mode buck controller. Figure 1 illustrates the block diagram for the control loop. The output voltage variation due to load or line changes will be sensed by the inverting input of the transconductance error amplifier via the feedback resistors (RFB1 and RFB2 in "Typical Application"), and compared to a reference voltage at the non-inverting input. This will cause a small change in the DC voltage level at the output of the error amplifier, or VCOMP. Meanwhile, the inductor current is sensed through the bottom MOSFET RDS(ON) and "Bottom Current Sense Circuit" as VIL. If VIL is lower than VCOMP, an ON-time period is triggered, in which DH pin is logic high and DL pin is logic low. The ON-time period length is predetermined by the "Fixed Ton Estimator" circuitry:
300kHz switching frequency. The actual ON-time varies a little with the different rising and falling times of the external MOSFETs. Therefore, the type of the external MOSFETs, the output load current, and the control circuitry power supply VIN will slightly modify the actual ON-time and the switching frequency. Also, the minimum TON, which is 140ns typical, results in a lower switching frequency in high VHSD and low VOUT applications, such as 18V to 0.8V. During the load transient, the switching frequency is changed due to the varying OFF-time. To illustrate the control loop, the steady-state scenario and the load transient scenario are analyzed. VCOMP is defined as the output of the error amplifier. Figure 2 shows the MIC2124 control loop timing during the steady-state operation in continuous mode. VIL represents the inductor current sensing voltage via the bottom MOSFET RDS(ON) and "Bottom Current Sense Circuit". When VIL is below VCOMP, which means that the inductor current reaches the valley value, the OFF-time ends and ON-time is triggered. The ON-time is predetermined by the estimation.
TON(estimated) =
VOUT VHSD 300kHz
(1)
where VOUT is the output voltage, VHSD is the power stage input voltage. After an ON-time period, the MIC2124 goes into the OFF-time period, in which DH pin is logic low and DL pin is logic high. The inductor current and VIL decrease during OFF time. If VIL is above VCOMP, the OFF status is maintained. When VIL is below VCOMP, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the inductor current and VCOMP is less than the minimum OFF time TOFF(min), which is 350ns typical, the MIC2124 control logic will apply the TOFF(min) instead. TOFF(min) is required to maintain enough energy in the Boost Capacitor (CBST) to drive the highside MOSFET. The maximum duty cycle is obtained from the 350ns TOFF(min):
DMAX = TS - TOFF(min) TS = 1- 350ns TS
Figure 2. MIC2124 Control Loop Timing
where TS = 1/300kHz = 3.33s. It is not recommended to use MIC2124 with a OFF-time close to TOFF(min) during steady state operation. The estimated ON-time method results in a constant
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Figure 3 shows the load transient operation of the MIC2124 converter. Assume the output voltage drops due to sudden load increase, which would cause the inverting input of the error amplifier, which is divided down version of VOUT, to be slightly less than the reference voltage, causing the output voltage of the error amplifier VCOMP to go high. This will cause "CONTROL LOGIC" to trigger ON-time period. At the end of the ONtime period, a minimum OFF-time TOFF(min) is generated to charge BST since the inductor current VIL is still below VCOMP. Then, the next ON-time period is triggered due to the high VCOMP. Therefore, the switching frequency changes during the load transient. Also the load regulation and transient load recovery is done by modulating the OFF-time. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in MIC2124 converter.
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Micrel, Inc.
MIC2124 until VLX > -127mV, and then goes into the ON status with minimum ON-time. The current limit threshold VCL has a fold back characteristic related to the FB voltage. Please refer to the "Typical Characteristics" for the curve of VCL vs. FB voltage. The circuit in Figure 4 illustrates the MIC2124 current limiting circuit.
Figure 3. MIC2124 Load-Transient Response Figure 4. MIC2124 Current Limiting Circuit
Unlike in current-mode control, the MIC2124 uses adaptive ON-time current mode control. The MIC2124 predetermined ON-time control loop has the advantage of constant ON-time mode control and eliminates the need for the slope compensation.
Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. The MIC2124 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 4ms. Therefore, the output voltage is controlled to increase slowly by a stair-case VREF ramp. Once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. VIN must be powered up no earlier than VHSD to make the soft-start function behavior correctly. Current Limit The MIC2174/MIC2174C uses the RSD(ON) of the lowside power MOSFET to sense over-current conditions. This method will avoid adding cost, board space and power losses taken by a discrete current sense resistor. The low-side MOSFET is used because it displays much lower parasitic oscillations during switching than the high-side MOSFET. In each switching cycle of the MIC2124 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. The sensed voltage VLX is compared with a current-limit threshold voltage VCL after a blanking time of 150ns. If the sensed voltage VLX is under VCL, which is -127mV typical at 0.8V feedback voltage, the MIC2124 keeps the low-side MOSFET on
Using the typical VCL value of -127mV, the current limit value in the inductor is roughly estimated as:
ICL 127mV R DS(ON)
For designs where the inductor current ripple is significant compared to the load current IOUT, or for low duty cycle operation, calculating the load current limit ICL(LOAD) should take into account that one is sensing the peak inductor current.
ICL(LOAD) = 127mV IL(pp) - RDS(ON) 2
(2) (3)
IL(pp) =
VOUT x (1- D) f SW xL
where: VOUT = The output voltage IL(pp) = Inductor current ripple peak-to-peak value D = Duty Cycle fSW = Switching frequency The MOSFET RDS(ON) varies 30% to 40% with temperature; therefore, it is recommended to add 50% margin to ICL(LOAD) in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect LX pin directly to the drain of the low-side MOSFET to accurately sense the MOSFETs RDS(ON)
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MOSFET Gate Drive The MIC2124 high-side drive circuit is designed to switch an N-Channel MOSFET. The typical application circuit shows a bootstrap circuit, consisting of a Schottky diode D1 and 0.1F boostrap capacitor CBST, as shown in the typical application schematic on Page 1. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the LX pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the highside MOSFET turns on, the voltage on the LX pin increases to approximately VHSD. Diode D1 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1F to 1F is sufficient to
MIC2124 hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e., BST = 10mA x 3.33s/0.1F = 333mV. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor RG, which is in series with CBST, can be used to slow down the turn-on time of the high-side N-Channel MOSFET. The drive voltage is derived from the supply voltage VIN. The nominal low-side gate drive voltage is VIN and the nominal high-side gate drive voltage is approximately VIN - VDIODE, where VDIODE is the voltage drop across D1. A dead-time of approximate 30ns between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs.
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MIC2124
Application Information
MOSFET Selection The MIC2124 controller works from input voltages of 3V to 18V and has an external 3V to 5.5V VIN supply to provide power to turn the external N-Channel power MOSFETs for the high-side and low-side switches. For applications where VIN < 5V, it is necessary that the power MOSFETs used are sub-logic level and are in full conduction mode for VGS of 2.5V. For applications when VIN > 5V; logic-level MOSFETs, whose operation is specified at VGS = 4.5V must be used. There are different criteria for choosing the high-side and low-side MOSFETs. These differences are more significant at lower duty cycles such as 12V to 1.8V conversion. In such an application, the high-side MOSFET is required to switch as quickly as possible to minimize transition losses, whereas the low-side MOSFET can switch slower, but must handle larger RMS currents. When the duty cycle approaches 50%, then the on-resistance of the high-side MOSFET starts to become critical. It is important to note that the on-resistance of a MOSFET increases with increasing temperature. A 75C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current limit. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2124 gate-drive circuit. At 300kHz switching frequency and above, the gate charge can be a significant source of power dissipation in the MIC2124. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is:
For the low-side MOSFET:
IG[low - side] (avg) = C ISS x VGS x f SW (5) Since the current from the gate drive comes from the VIN, the power dissipated in the MIC2124 due to gate drive is: PGATEDRIVE = VIN .(IG[high - side] (avg) + IG[low -side] (avg)) (6) A convenient figure of merit for switching MOSFETs is the on-resistance times the total gate charge RDS(ON) x QG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2124. Also, the RDS(ON) of the low-side MOSFET will determine the current limit value. Please refer to "Current Limit" subsection in "Functional Description" for more details. Parameters that are important to MOSFET switch selection are: * * * Voltage rating On-resistance
Total gate charge The voltage ratings for the high-side and low-side MOSFETs are essentially equal to the power stage input voltage VHSD. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitic elements. The power dissipated in the MOSFETs is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses during the period of time when the MOSFETs turn on and off (PAC). PSW = PCONDUCTION + PAC (7)
PCONDUCTION = ISW(RMS) x R DS(ON) PAC = PAC(off ) + PAC(on)
2
(8) (9)
IG[high-side] (avg) = Q G x f SW
(4)
where: IG[high-side](avg) = Average high-side MOSFET gate current QG = Total gate charge for the high-side MOSFET taken from the manufacturer's data sheet for VGS = VIN. fSW = Switching Frequency (300kHz) The low-side MOSFET is turned on and off at VDS = 0 because an internal body diode or external freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge.
where: RDS(ON) = on-resistance of the MOSFET switch D = Duty Cycle = VOUT / VHSD Making the assumption that the turn-on and turn-off transition times are equal; the transition times can be approximated by: tT = C ISS x VIN + C OSS x VHSD IG (10)
where: CISS and COSS are measured at VDS = 0 IG = gate-drive current The total high-side MOSFET switching loss is: PAC = (VHSD + VD ) x IPK x t T x f SW (11)
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Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below.
MIC2124
Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below:
PINDUCTORCu = IL(RMS) R WINDING
2
(16)
The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. R WINDING(Ht) = R WINDING(20 C) (1 + 0.0042 (TH - T20C )) (17) where: TH = temperature of wire under full load T20C = ambient temperature RWINDING(20C) = room temperature winding resistance (usually specified by the manufacturer)
Output Capacitor Selection The type of the output capacitor is usually determined by its ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitors are tantalum, low-ESR aluminum electrolytic, OS-CON and POSCAPS. The output capacitor's ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. See "Feedback Loop Compensation" section for more information. The maximum value of ESR is calculated:
L=
VOUT VHSD(max) - VOUT
(
)
VHSD f SW 20% I OUT(max)
(12)
where: fSW = switching frequency, 300 kHz 20% = ratio of AC ripple current to DC output current VHSD(max) = maximum power stage input voltage The peak-to-peak inductor current ripple is: IL(PP) = VOUT (VHSD(max) - VOUT ) VHSD(max) f SW L (13)
The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(pk) = I OUT(max) + 0.5 x IL(PP) (14)
2
The RMS inductor current is used to calculate the I R losses in the inductor.
ESR COUT
VOUT(PP) IL(PP)
(18)
IL(RMS) = I OUT(max) +
2
IL(PP) 12
2
(15)
Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2124 requires the use of ferrite materials for all but the most cost sensitive applications.
where: VOUT(PP) = peak-to-peak output voltage ripple IL(PP) = peak-to-peak inductor current ripple The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated below:
VOUT(PP) IL(PP) 2 + IL(PP) ESR C = OUT C f SW 8 OUT (19)
2
(
)
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2
MIC2124 R2 = VREF R1 VOUT - VREF (26)
(20)
The power dissipated in the output capacitor is: PDISS(COUT ) = I COUT (RMS) ESR COUT (21)
Figure 5. Voltage-Divider Configuration
Input Capacitor Selection The input capacitor for the power stage input VHSD should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor's voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. The input voltage ripple will primarily depend on the input capacitor's ESR. The peak input current is equal to the peak inductor current, so:
External Schottky Diode (Optional) An external freewheeling diode, which is not necessary, is used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead-time prevents current from flowing unimpeded through both MOSFETs and is typically 30ns. The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current.
ID(avg) = IOUT 2 30ns f SW The reverse voltage requirement of the diode is: VDIODE(rrm) = VHSD The power dissipated by the Schottky diode is: PDIODE = ID(avg) x VF
(27)
VIN = IL(pk) ESR CIN
(22)
The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: ICIN (RMS) IOUT(MAX) D (1 - D) The power dissipated in the input capacitor is: PDISS(CIN ) = I CIN (RMS) ESR CIN
2
(28)
(23)
(24)
Voltage Setting Components The MIC2124 requires two resistors to set the output voltage as shown in Figure 5. The output voltage is determined by the equation:
R1 ) (25) R2 where VREF = 0.8V. A typical value of R1 can be between 3k and 10k. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: VOUT = VREF (1 +
where VF = forward voltage at the peak diode current. The external Schottky diode is not necessary for the circuit operation since the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease the high frequency noise. If the MOSFET body diode is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the circuit inductance will cause ringing during the high-side MOSFET turn-on. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending on the circuit components and operating conditions, an external Schottky diode will give a 0.5% to 1% improvement in efficiency. 14
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Feedback Loop Compensation The MIC2124 controller comes with an internal error amplifier used for optimizing control loop stability by placing a capacitor C1 in series with a resistor R1 and another capacitor C2 in parallel from the COMP pin to ground.
MIC2124
and the output inductor in the power stage: f z( con) = 1 2 x C OUT x ESR COUT (30)
fp(con) =
1 1 1 D x( + x) 2 C OUT x R LOAD f SW x L x C OUT 2
(31)
Therefore, type II compensation, which is comprised by C1, R1 and C2, is able to achieve a stabilized loop for MIC2124 in most applications.
b. gm Error Amplifier It is undesirable to have high error amplifier gain at high frequencies because high frequency noise spikes would be picked up and transmitted at large amplitude to the output; thus, gain should be permitted to fall off at high frequencies. At low frequency, it is desired to have high open-loop gain to attenuate the power line ripple. Thus, the error amplifier gain should be allowed to increase rapidly at low frequencies. The transfer function with R1, C1, and C2 for the internal gm error amplifier can be expressed as:
1 + s x R1 x C1 (32) = gm x C1 x C2 s x (C1 + C2) x 1 + s x R1 x C1 + C2
Figure 6. Loop Compensation
a. Power Stage The adaptive on-time current mode control applied in MIC2124 controller eliminates the double-pole in the power stage, which is caused by the output inductor and output capacitor. At the frequency range which is far below the switching frequency (f < fSW /6), the transfer function from the output of the error amplifier to the buck converter output can be approximated by the following equation:
G(s)err
One pole and one zero can be seen from the above transfer function at the following frequencies:
f z( err ) = f p( err ) = 1 2 x R1 x C1 1 2 x R1 x C1 x C2 C1 + C2 (33) (34)
G(s) con G C x
1 + s x C OUT x ESR COUT s 1+ p
(29)
where:
GC =
R LOAD x Ri
1 1+ R LOAD D x f SW x L 2
p =
1 1 D + x C OUT x R LOAD f SW x L x C OUT 2
c. Total Open-Loop Response The open-loop response for the MIC2124 controller is easily obtained by combining the power stage, the feedback resistor divider, and the error amplifier gains together.
COUT = total output capacitors ESRCOUT = electrical series resistance of the output capacitor RLOAD = load resistance Ri = 2.4 x Rds(on)_bottom (low-side MOSFET Rds(on)) fSW = switching frequency L = inductance of the output inductor D = duty cycle According to equation (29), there is a pole and zero pair set by the load resistance RLOAD, the output capacitor,
G(s) total =
R FB2 x G(s) con x G(s) err (35) R FB1 + R FB2
where RFB1 and RFB2 are the voltage divider resistors, as shown in the typical application schematic on Page 1. It is desirable to have the gain curve intersect zero dB at tens of kilohertz, this is commonly called crossover frequency; the phase margin at crossover frequency should be at least 45. 12V to 1.8V @ 10A application is applied as an example to demonstrate the loop compensation for MIC2124. In this application:
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Micrel, Inc. D = 0.15 RLOAD = 0.18. The output capacitor and the inductor parameters are: COUT = 760F ESRCOUT = 0.002 L = 2.2H. Also, Ri = 0.007x2.4 fSW = 300kHz Rfb1 = 10k Rfb2 = 8.06k The error amplifier gm and external compensation component are: gm = 110S R1 = 150k C1 = 220pF C2 = 47pF The gain and phase of the control-to-output transfer function predicted by the equation (29) are shown in Figure 7. The gain and phase of the error amplifier transfer function predicted by the equation (32) are shown in Figure 8. The total open-loop bode plot predicted by the equation (35) is shown in Figure 9.
30 0
MIC2124
90 Gain of Error Amplifier Phase of Error Amplifier
20
70
0
30
-40
10
-60
-10
-80
-30 10 100 1000 10000
-100 100000
f (Hz)
Figure 8. Error Amplifier Bode Plot
120 Total Open Loop Gain Total Open Loop Phase
0
90
-30
30
-90
0
-120
-30
20 -20
-150
-60
Phase (Degree)
10
-40
10
100
1000
10000
-180 100000
Gain (dB)
f (Hz)
0
-60
-10
-80
Figure 9. Total Open Loop Bode Plot
-20
Gain of Control-to-Output Phase of Control-to-Output 10 100 1000 10000
-100
-30
-120 100000
The crossover frequency of this MIC2124 buck converter is 40kHz and the phase margin is about 50, as shown in Figure 9.
f (Hz)
Figure 7. Control-to-Output Bode Plot
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Phase (Degree)
60
-60
Gain (dB)
Phase (Degree)
50
-20
Gain (dB)
Micrel, Inc.
MIC2124
Inductor
PCB Layout Guideline
Warning!!! To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC2124 converter. IC
* * * *
Keep the inductor connection to the switch node (LX) short. Do not route any digital lines underneath or close to the inductor. Keep the switch node (LX) away from the feedback (FB) pin. The LX pin should be connected directly to the drain of the low-side MOSFET to accurate sense the voltage across the low-side MOSFET.
*
The 2.2F ceramic capacitor, which connects to the VIN terminal, must be located right at the IC. The VIN terminal is very noise sensitive and placement of the capacitor is very critical. Use wide traces to connect to the IN and PGND pins. Place the IC and MOSFETs close to the point of load (POL). Use fat traces to route the input and output power lines.
*
* * *
To minimize noise, place a ground plane underneath the inductor. Output Capacitor * Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. * Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM.
Signal and power grounds should be kept separate and connected at only one location. Input Capacitor * Place the HSD input capacitor next. * Place the HSD input capacitors on the same side of the board and as close to the MOSFETs and the IC as possible. Keep both the HSD and PGND connections short. Place several vias to the ground plane close to the HSD input capacitor ground terminal. Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be de-rated by 50%. In "Hot-Plug" applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. An additional Tantalum or Electrolytic bypass input capacitor of 22uF or higher is required at the input power connection. The 2.2F, which connect to the VIN terminal, must be located right at the IC. The VIN terminal is very noise sensitive and placement of the capacitor is very critical. Connections must be made with wide trace. June 2010 17
*
* * * *
The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation. Schottky Diode (Optional) * Place the Schottky diode on the same side of the board as the MOSFETs and HSD input capacitor. * The connection from the Schottky diode's Anode to the input capacitors ground terminal must be as short as possible.
*
*
*
The diode's Cathode connection to the switch node (LX) must be kept as short as possible. RC Snubber * Place the RC snubber on the same side of the board and as close to the MOSFETs as possible. MOSFETS * Low-side MOSFET gate drive trace (DL pin to MOSFET gate pin) must be short and routed over a ground plane. The ground plane should be the connection between the MOSFET source and PGND. * Chose a low-side MOSFET with a high CGS/CGD ratio and a low internal gate resistance to minimize the effect of dv/dt inducted turn-on. Do not put a resistor between the LSD output and the gate. Use a 4.5V VGS rated MOSFET. Its higher gate threshold voltage is more immune to glitches than a 2.5V or 3.3V rated MOSFET. MOSFETs that are rated for operation at less than 4.5V VGS should not be used.
M9999-060810-D
*
* *
*
Micrel, Inc.
Others
MIC2124 * The compensation resistor and capacitors should be placed right next to the COMP pin and the other side should connect directly to the GND pin on the MIC2124 rather than going to the plane. HSD pin is sensitive to the noise. Too much noise at HSD pin may cause the jittering at LX. A 10 resistor and 0.1F capacitor low-pass filter at the HSD is able to mitigate the noise.
*
In order to accurately sense the voltage across the low-side MOSFET, the LX pin and PGND pin should be Kelvin connected to the drain and source of the low-side MOSFET. The feedback resistors RFB1 and RFB2 (refer to the typical application schematic on page 1) should be placed close to the FB pin. The top side of RFB1 should connect directly to the output node. Run this trace away from the switch node (junction of Q1, Q2, and the output inductor).
*
*
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MIC2124
Evaluation Board Schematic
Figure 10. Schematic of MIC2124 5A Evaluation Board
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MIC2124
Bill of Materials
Item Part Name Manufacturer Description Qty
C1 C2
B41125A7227M 1210YD226MAT2A GRM32ER61C226ME20L C3225XR1C226M
EPCOS AVX TDK Murata
(1)
220F Aluminum Capacitor, SMD 35V 22F Ceramic Capacitor, X5R, Size 1210, 16V
1 1
(2) (3)
(4) (5)
C3 C6, C9, C13
EEFSX0D181R 06035C10KAT2A GRM21BR71A225KA01L C1608X7R1H104K 0805ZC225MAT2A GRM21BR71A225KA01L C2012X7R1A225K
Panasonic AVX Murata TDK AVX Murata TDK AVX Murata TDK AVX Murata TDK AVX Murata TDK MCC
180F SP Capacitor, 9m, 2V 0.1F Ceramic Capactior, X7R, Size 0603, 50V
1 3
C7, C8
2.2F Ceramic Capacitor, X7R, Size 0805, 10V
2
C10
06035C471KAT2A GRM188R71H471KA01D C1608X7R1H471K
470pF Ceramic Capacitor, X7R, Size 0603, 50V
1
C11
06035C102KAT2A GRM188R71H102KA01D C1608X7R1H102K
1nF Ceramic Capacitor, X7R, Size 0603, 50V
1
C12
06035A180JAT2A GRM1885C1H180JA01D C1608C0G1H180J
18pF Ceramic Capacitor, Size 0603, 50V
1
D1 L1 Q1 R1, R5 R2 R3 R4 R6 R7 U1 U2
(10)
SD103BWS SD103BWS-7 DO316P-272HC FDS6890A CRCW06030000Z0EA CRCW08051R21FKEA CRCW060382K0FKEA CRCW06037K15FKEA CRCW06038K06FKEA
MIC2124YMM MIC5233-5.0YM5
30V small signal Schottky diode
(6)
1 1 1 2 1 1 1 1 1 1
DIODE INC Coilcraft Fairchild
2.7H inductor, 6.6A saturation current 20V, 7.5A Dual N-MOSFET, 0.018 Rds (on) @ 4.5V 0 resistor, size 0603, 1% 1.21 resistor, size 0805, 1% 82k resistor, size 0603, 1% 7.15k resistor, size 0603, 1% 8.06k resistor, size 0603, 1% Open 300kHz Buck Controller LDO
(8)
(7)
Vishay/Dale
Vishay/Dale Vishay/Dale Vishay/Dale Vishay/Dale
Micrel, Inc
(9)
Micrel, Inc
Notes: 1. EPCOS: www.epcos.com 2. AVX: www.avx.com 3. Murata: www.murata.com 4. TDK: www.tdk.com 5. Panasonic: www.panasonic.com 6. Diodes Inc.: www.diodes.com 7. Fairchild: www.fairchildsemi.com 8. Vishay: www.vishay.com 9. Micrel, Inc.: www.micrel.com 10. Optional: Required if 5V supply is not available in the system.
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MIC2124
PCB Layout
Figure 11. MIC2124 Evaluation Board Top layer
Figure 12. MIC2124 Evaluation Board Mid-Layer 1(Ground Plane)
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MIC2124
Figure 13. MIC2124 Evaluation Board Mid Layer 2
Figure 14. MIC2124 Evaluation Board Bottom Layer
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MIC2124
Package Information
10-Pin MSOP (MM)
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MIC2124
Recommended Landing Pattern
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2010 Micrel, Incorporated.
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